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Less expensive than many mechanical switches Projects a `touch button' through any dielectric 100% autocal for life - no adjustments required No active external components Piezo sounder direct drive for `tactile' click feedback LED drive for visual feedback 2.5 ~ 5V single supply operation 10A at 2.5V - very low power drain Toggle mode for on/off control (via option pins) 10s or 60s auto-recalibration timeout (via option pins) Pulse output mode (via option pins) Gain settings in 3 discrete levels Simple 2-wire operation possible HeartBeatTM health indicator on output Active low (QT110), active high (QT110H) versions
QT110 / QT110H
QTOUCHTM SENSOR ICs
Vdd Out Opt1 Opt2
1
8
Vss Sns2 Sns1 Gain
QT110
2 3 4
7 6 5
APPLICATIONS Light switches Industrial panels Appliance control Security systems Access systems Pointing devices Elevator buttons Consumer electronics
The QT110 / QT110H charge-transfer ("QT'") sensor chips are self-contained digital ICs used to implement near-proximity or touch sensors. They project sense fields through almost any dielectric, like glass, plastic, stone, ceramic, and wood. They can also turn small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with an ability to self-calibrate continuously leads to entirely new product concepts. These devices are designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical switch or button may be found; they may also be used for some material sensing and control applications provided that the presence duration of objects does not exceed the recalibration timeout interval. A piezo element can also be connected to create a feedback click sound. These ICs require only a common inexpensive capacitor in order to function. Average power consumption is under 20A in most applications, allowing battery operation. The devices employ digital signal processing techniques pioneered by Quantum, designed to make them survive real-world challenges, such as `stuck sensor' conditions and signal drift. Sensitivity is digitally determined for the highest possible stability. No external active components are required for operation. The devices include several user-selectable built-in features. One, toggle mode, permits on/off touch control for example for light switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power rail, permitting a simple 2-wire twisted-pair interface. Quantum's unique HeartBeatTM signal is also included, allowing a host controller to continuously monitor sensor health. By using the charge transfer principle, these ICs deliver a level of performance clearly superior to older technologies in a highly cost-effective package.
TA
00C to +700C 00C to +700C -400C to +850C -400C to +850C
AVAILABLE OPTIONS SOIC
QT110-IS QT110H-IS
8-PIN DIP
QT110-D QT110H-D -
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(c)1999-2004 Quantum Research Group
QT110/110H R1.03/0604
1 - OVERVIEW
The QT110 is a digital burst mode charge-transfer (QT) sensor designed specifically for touch controls; it includes all hardware and signal processing functions necessary to provide stable sensing under a wide variety of changing conditions. Only a few low cost, non-critical discrete external parts are required for operation.
Figure 1-1 Standard mode options
+2.5 ~ +5 1 2 3 4
OUTPUT = DC TIMEOUT = 10 Secs TOGGLE = OFF GAIN = HIGH Vdd OUT SNS2
Figure 1-1 shows the basic QT110 circuit using the device, with a conventional output drive and power supply connections. Figure 1-2 shows a second configuration using a common power/signal rail which can be a long twisted pair from a controller; this configuration uses the built-in pulse mode to transmit output state to the host controller (QT110 only).
7 5 Cs 6
2nF - 500nF
RE
SENSING ELECTRODE
OPT1
GAIN
Rs
Cx
OPT2 Vss
SNS1
1.1 BASIC OPERATION
The QT110 employs low duty cycle bursts of charge-transfer cycles to acquire its signal. Burst mode permits power consumption in the low microamp range, dramatically reduces EMC problems, and yet permits excellent response time. Internally the signals are digitally processed to reject impulse noise, using a 'consensus' filter which requires four consecutive confirmations of a detection before the output is activated. The QT switches and charge measurement hardware functions are all internal to the QT110 (Figure 1-3). A single-slope switched capacitor ADC includes both the required QT charge and transfer switches in a configuration that provides direct ADC conversion. Vdd is used as the charge reference voltage. Larger values of Cx cause the charge transferred into Cs to rise more rapidly, reducing available resolution; as a minimum resolution is required for proper operation, this can result in dramatically reduced apparent gain. The IC is highly tolerant of changes in Cs since it computes the signal threshold level ratiometrically. Cs is thus non-critical and can be an X7R type. As Cs changes with temperature, the internal drift compensation mechanism also adjusts for the drift automatically. Piezo sounder drive: The QT110 can drive a piezo sounder after a detection for feedback. The piezo sounder replaces or augments the Cs capacitor; this works since piezo sounders are also capacitors, albeit with a large thermal drift coefficient. If Cpiezo is in the proper range, no additional capacitor is required. If Cpiezo is too small, it can simply be `topped up' with a ceramic capacitor in parallel. The QT110 drives a ~4kHz signal across SNS1 and SNS2 to make the piezo (if installed) sound a short tone for 75ms immediately after detection, to act as an audible confirmation. Option pins allow the selection or alteration of several other special features and sensitivity.
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1.2 ELECTRODE DRIVE
The internal ADC treats Cs as a floating transfer capacitor; as a direct result, the sense electrode can in theory be connected to either SNS1 or SNS2 with no performance difference. However, the noise immunity of the device is improved by connecting the electrode to SNS2, preferably via a series resistor Re (Figure 1-1) to roll off higher harmonic frequencies, both outbound and inbound. In order to reduce power consumption and to assist in discharging Cs between acquisition bursts, a 470K series resistor Rs should be connected across Cs (Figure 1-1). The rule Cs >> Cx must be observed for proper operation. Normally Cx is on the order of 10pF or so, while Cs might be 10nF (10,000pF), or a ratio of about 1:1000. It is important to minimize the amount of unnecessary stray capacitance Cx, for example by minimizing trace lengths and widths and backing off adjacent ground traces and planes so as keep gain high for a given value of Cs, and to allow for a larger sensing electrode size if so desired. The PCB traces, wiring, and any components associated with or in contact with SNS1 and SNS2 will become touch sensitive and should be treated with caution to limit the touch area to the desired location.
1.3 ELECTRODE DESIGN
1.3.1 ELECTRODE GEOMETRY AND SIZE
Figure 1-2 2-wire operation, self-powered
+
3.5 - 5.5V CMOS LOGIC 1K Twisted pair
There is no restriction on the shape of the electrode; in most cases common sense and a little experimentation can result in a good electrode design. The QT110 will operate equally well with long, thin electrodes as with round or square ones; even random shapes are acceptable. The electrode can also be a 3-dimensional surface or object. Sensitivity is related to electrode surface area, orientation with respect to the object being sensed, object composition, and the ground coupling quality of both the (not QT110H) sensor circuit and the sensed object.
1.3.2 KIRCHOFF'S CURRENT LAW
10F
1N4148
1 2
Vdd SNS2 OUT
7 5 6 Cs Rs
RE
SENSING ELECTRODE
n-ch Mosfet
3 4
OPT1
GAIN
Cx
OPT2
SNS1
Vss
8
Like all capacitance sensors, the QT110 relies on Kirchoff's Current Law (Figure 1-5) to detect the change in capacitance of the electrode. This law as applied to capacitive sensing requires that the sensor's field current must complete a loop, returning back to its source in order for capacitance to be sensed. Although most designers relate to Kirchoff's law with regard to hardwired circuits, it applies equally to capacitive
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field flows. By implication it requires that the signal ground and the target object must both be coupled together in some manner for a capacitive sensor to operate properly. Note that there is no need to provide actual hardwired ground connections; capacitive coupling to ground (Cx1) is always sufficient, even if the coupling might seem very tenuous. For example, powering the sensor via an isolated transformer will provide ample ground coupling, since there is capacitance between the windings and/or the transformer core, and from the power wiring itself directly to 'local earth'. Even when battery powered, just the physical size of the PCB and the object into which the electronics is embedded will generally be enough to couple a few picofarads back to local earth.
Figure 1-3 Internal Switching & Timing
Result Single -Slo pe 14-bit Switched Capacitor ADC
E LEC TRO DE
S NS2
Bu rst Controller
Cs Cx
S NS1
Start
Do ne
C ha rg e Amp
1.3.3 VIRTUAL CAPACITIVE GROUNDS
When detecting human contact (e.g. a fingertip), grounding of the person is never required. The human body naturally has several hundred picofarads of `free space' capacitance to the local environment (Cx3 in Figure 1-3), which is more than two orders of magnitude greater than that required to create a return path to the QT110 via earth. The QT110's PCB however can be physically quite small, so there may be little `free space' coupling (Cx1 in Figure 1-3) between it and the environment to complete the return path. If the QT110 circuit ground cannot be earth grounded by wire, for example via the supply connections, then a `virtual capacitive ground' may be required to increase return coupling. A `virtual capacitive ground' can be created by connecting the QT110's own circuit ground to: - A nearby piece of metal or metallized housing; - A floating conductive ground plane; - Another electronic device (to which its might be connected already). Free-floating ground planes such as metal foils should maximize exposed surface area in a flat plane if possible. A square of metal foil will have little effect if it is rolled up or crumpled into a ball. Virtual ground planes are more effective and can be made smaller if they are physically bonded to other surfaces, for example a wall or floor.
In some cases it may be desirable to increase sensitivity further, for example when using the sensor with very thick panels having a low dielectric constant. Sensitivity can often be increased by using a bigger electrode, reducing panel thickness, or altering panel composition to one having a higher dielectric constant. Increasing electrode size can have diminishing returns, as high values of Cx will reduce sensor gain. Increasing the electrode's surface area will not substantially increase touch sensitivity if its diameter is already much larger in surface area than the object being detected. Metal areas near the electrode will reduce the field strength and increase Cx loading and are to be avoided for maximal gain. Ground planes around and under the electrode and its SNS trace will cause high Cx loading and destroy gain. The possible signal-to-noise ratio benefits of ground area are more than negated by the decreased gain from the circuit, and so ground areas around electrodes are discouraged. Keep ground, power, and other signals traces away from the electrodes and SNS wiring. The value of Cs has a minimal effect on sensitivity with these devices, but if the Cs value is too low there can be a sharp drop-off in sensitivity.
1.3.4 SENSITIVITY
The QT110 can be set for one of 3 gain levels using option pin 5 (Table 1-1). If left open, the gain setting is high. The sensitivity change is made by altering the numerical threshold level required for a detection. It is also a function of other things: electrode size, shape, and orientation, the composition and aspect of the object to be sensed, the thickness and composition of any overlaying panel material, and the degree of ground coupling of both sensor and object are all influences. Gain plots of the device are shown on page 9. The Gain input should never be tied to anything other than SNS1 or SNS2, or left unconnected (for high gain setting).
Se nse E le ctro de
Figure 1-5 Kirchoff's Current Law
CX2
SENSO R
Table 1-1 Gain Strap Options
Gain High Medium Low Tie Pin 5 to: Leave open Pin 6 Pin 7
CX 1 CX 3
Su rro und ing e nv iro nm en t
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2 - QT110 SPECIFICS
2.1 SIGNAL PROCESSING
The QT110 processes all signals using a number of algorithms pioneered by Quantum. The algorithms are specifically designed to provide for high 'survivability' in the face of all kinds of adverse environmental changes.
2.1.4 DETECTION INTEGRATOR
It is desirable to suppress detections generated by electrical noise or from quick brushes with an object. To accomplish this, the QT110 incorporates a detect integration counter that increments with each detection until a limit is reached, after which the output is activated. If no detection is sensed prior to the final count, the counter is reset immediately to zero. In the QT110, the required count is 4. The Detection Integrator can also be viewed as a 'consensus' filter, that requires four detections in four successive bursts to create an output. As the basic burst spacing is 75ms, if this spacing was maintained throughout all 4 counts the sensor would react very slowly. In the QT110, after an initial detection is sensed, the remaining three bursts are spaced about 20ms apart, so that the slowest reaction time possible is 75+20+20+20 or 135ms and the fastest possible is 60ms, depending on where in the initial burst interval the contact first occurred. The response time will thus average about 95ms.
2.1.1 DRIFT COMPENSATION ALGORITHM
Signal drift can occur because of changes in Cx and Cs over time. It is crucial that drift be compensated for, otherwise false detections, non-detections, and sensitivity shifts will follow. Cs drift has almost no effect on gain since the threshold method used is ratiometric. However Cs drift can still cause false detections if the drift occurs rapidly. Drift compensation (Figure 2-1) is performed by making the reference level track the raw signal at a slow rate, but only while there is no detection in effect. The rate of adjustment must be performed slowly, otherwise legitimate detections could be ignored. The QT110 drift compensates using a slew-rate limited change to the reference level; the threshold and hysteresis values are slaved to this reference. Once an object is sensed, the drift compensation mechanism ceases since the signal is legitimately high, and therefore should not cause the reference level to change. The QT110's drift compensation is 'asymmetric': the reference level drift-compensates in one direction faster than it does in the other. Specifically, it compensates faster for decreasing signals than for increasing signals. Increasing signals should not be compensated for quickly, since an approaching finger could be compensated for partially or entirely before even touching the sense pad. However, an obstruction over the sense pad, for which the sensor has already made full allowance for, could suddenly be removed leaving the sensor with an artificially elevated reference level and thus become insensitive to touch. In this latter case, the sensor will compensate for the object's removal very quickly, usually in only a few seconds.
2.1.5 FORCED SENSOR RECALIBRATION
The QT110 has no recalibration pin; a forced recalibration is accomplished only when the device is powered up. However, the supply drain is so low it is a simple matter to treat the entire IC as a controllable load; simply driving the QT110's Vdd pin directly from another logic gate or a microprocessor port (Figure 2-2) will serve as both power and 'forced recal'. The source resistance of most CMOS gates and microprocessors is low enough to provide direct power without any problems. Almost any CMOS logic gate can directly power the QT110. A 0.01uF minimum bypass capacitor close to the device is essential; without it the device can break into high frequency oscillation. Option strap configurations are read by the QT110 only on powerup. Configurations can only be changed by powering the QT110 down and back up again; again, a microcontroller can directly alter most of the configurations and cycle power to put them in effect.
2.1.2 THRESHOLD CALCULATION
Sensitivity is dependent on the threshold level as well as ADC gain; threshold in turn is based on the internal signal reference level plus a small differential value. The threshold value is established as a percentage of the absolute signal level. Thus, sensitivity remains constant even if Cs is altered dramatically, so long as electrode coupling to the user remains constant. Furthermore, as Cx and Cs drift, the threshold level is automatically recomputed in real time so that it is never in error. The QT110 employs a hysteresis dropout below the threshold level of 50% of the delta between the reference and threshold levels. The threshold setting is determined by option jumper; see Section 1.3.4.
2.2 OUTPUT FEATURES
The devices are designed for maximum flexibility and can accommodate most popular sensing requirements. These are selectable using strap options on pins OPT1 and OPT2. All options are shown in Table 2-1. OPT1 and OPT2 should never be left floating. If they are floated, the device will draw excess power and the options will not be properly read on powerup. Intentionally, there are no pullup resistors on these lines, since pullup resistors add to power drain if the pin(s) are tied low.
2.2.1 DC MODE OUTPUT
The output of the device can respond in a DC mode, where the output is active-low (QT110) or active-high (QT110H) upon detection. The output will remain active for the duration of the
2.1.3 MAX ON-DURATION
If an object or material obstructs the sense pad the signal may rise enough to create a detection, preventing further operation. To prevent this, the sensor includes a timer which monitors detections. If a detection exceeds the timer setting, the timer causes the sensor to perform a full recalibration. This is known as the Max On-Duration feature. After the Max On-Duration interval, the sensor will once again function normally, even if partially or fully obstructed, to the best of its ability given electrode conditions. There are two nominal timeout durations available via strap option: 10 and 60 seconds. The accuracy of these timeouts is approximate.
Figure 2-1 Drift Compensation
Signal H ysteresis
Threshold R eference
Output
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Figure 2-2 Powering From a CMOS Port Pin
P O RT X .m 0.01F
Figure 2-3 Damping Piezo Clicks with Rs
+2.5 ~ +5 1 2
Vdd OUT SNS1
C MO S m icro controller
V dd P O RT X .n O UT
7 5 6
Piezo Sounder 10-30nF
RE
SENSING ELECTRODE
Q T110
V ss
3 4
OPT1
GAIN
Rs
Cx
OPT2 Vss
SNS2
8
detection, or until the Max On-Duration expires, whichever occurs first. If the latter occurs first, the sensor performs a full recalibration and the output becomes inactive until the next detection. In this mode, two Max On-Duration timeouts are available: 10 and 60 seconds.
should be added to bring the total Cs across SNS1 and SNS2 to at least 10nF, or possibly more if Cx is above 5pF Piezo sounders have very high, uncharacterized thermal coefficients and should not be used if fast temperature swings are anticipated, especially at high gains. They are also generally unstable at high gains; even if the total value of Cs is largely from an added capacitor the piezo can cause periodic false detections. The burst acquisition process induces a small but audible voltage step across the piezo resonator, which occurs when SNS1 and SNS2 rapidly discharge residual voltage stored on the resonator. The resulting slight clicking sound can be greatly reduced by placing a 470K resistor Rs in parallel with the resonator; this acts to slowly discharge the resonator, attenuating of the harmonic-rich audible step (Figure 2-3). Note that the piezo drive does not operate in Pulse mode.
2.2.2 TOGGLE MODE OUTPUT
This makes the sensor respond in an on/off mode like a flip flop. It is most useful for controlling power loads, for example in kitchen appliances, power tools, light switches, etc. Max On-Duration in Toggle mode is fixed at 10 seconds. When a timeout occurs, the sensor recalibrates but leaves the output state unchanged.
Table 2-1 Output Mode Strap Options
Tie Pin 3 to: DC Out DC Out Toggle Pulse Vdd Vdd Gnd Gnd Tie Pin 4 to: Vdd Gnd Gnd Vdd Max OnDuration 10s 60s 10s 10s
2.2.5 HEARTBEATTM OUTPUT
The output has a full-time HeartBeatTM `health' indicator superimposed on it. This operates by taking 'Out' into a 3-state mode for 350s once before every QT burst. This output state can be used to determine that the sensor is operating properly, or, it can be ignored using one of several simple methods. QT110: The HeartBeat indicator can be sampled by using a pulldown resistor on Out, and feeding the resulting negative-going pulse into a counter, flip flop, one-shot, or other circuit. Since Out is normally high, a pulldown resistor will create negative HeartBeat pulses (Figure 2-4) when the sensor is not detecting an object; when detecting an object, the output will remain active for the duration of the detection, and no HeartBeat pulse will be evident. QT110H: Same as QT110 but inverted logic (use a pull-up resistor instead of a pull-down). If the sensor is wired to a microcontroller as shown in Figure 2-5, the controller can reconfigure the load resistor to either ground or Vcc depending on the output state of the device, so that the pulses are evident in either state. Electromechanical devices will usually ignore this short pulse. The pulse also has too low a duty cycle to visibly activate LED's. It can be filtered completely if desired, by adding an RC timeconstant to filter the output, or if interfacing directly and only to a high-impedance CMOS input, by doing nothing or at most adding a small non-critical capacitor from Out to ground (Figure 2-6).
2.2.3 PULSE MODE OUTPUT
This mode generates a pulse of 75ms duration (QT110 negative-going; QT110H - positive-going) with every new detection. It is most useful for 2-wire operation, but can also be used when bussing together several devices onto a common output line with the help of steering diodes or logic gates, in order to control a common load from several places. Max On-Duration is fixed at 10 seconds if in Pulse output mode. Note that the beeper drive does not operate in Pulse mode.
2.2.4 PIEZO ACOUSTIC DRIVE
A piezo drive signal is generated for use with a piezo sounder immediately after a detection is made; the tone lasts for a nominal 95ms to create a `tactile feedback' sound. The sensor drives the piezo using an H-bridge configuration for the highest possible sound level. The piezo is connected across pins SNS1 and SNS2 in place of Cs or in addition to a parallel Cs capacitor. The piezo sounder should be selected to have a peak acoustic output in the 3.5kHz to 4.5kHz region. Since piezo sounders are merely high-K ceramic capacitors, the sounder will double as the Cs capacitor, and the piezo's metal disc can even act as the sensing electrode. Piezo transducer capacitances typically range from 6nF to 30nF in value; at the lower end of this range an additional capacitor
2.2.6 OUTPUT DRIVE
The QT110's output is active low (QT110) or active high (QT110H) and can source 1mA or sink 5mA of non-inductive current.
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Figure 2-4
Getting HB pulses with a pull-down resistor (QT110 shown; use pull-up resistor with QT110H)
Figure 2-5
Using a micro to obtain HB pulses in either output state (QT110 or QT110H)
+2 .5 to 5
H eartBeatTM P ulses
1 2
Vdd O UT S NS 2
7 5 6
M icro processo r
P O RT _ M .x
2
Ro
OU T
SN S 2
7 5 6
Ro
3 4
O PT 1
GAIN
3 4
OP T 1
GA IN
O PT 2 Vss
S NS 1
P O RT _ M .y
OP T 2
SN S 1
8
Care should be taken when the IC and the load are both powered from the same supply, and the supply is minimally regulated. The device derives its internal references from the power supply, and sensitivity shifts can occur with changes in Vdd, as happens when loads are switched on. This can induce detection `cycling', whereby an object is detected, the load is turned on, the supply sags, the detection is no longer sensed, the load is turned off, the supply rises and the object is reacquired, ad infinitum. To prevent this occurrence, the output should only be lightly loaded if the device is operated from an unregulated supply, e.g. batteries. Detection `stiction', the opposite effect, can occur if a load is shed when Out is active. QT110: The output of the QT110 can directly drive a resistively limited LED. The LED should be connected with its cathode to the output and its anode towards Vcc, so that it lights when the sensor is active-low. If desired the LED can be connected from Out to ground, and driven on when the sensor is inactive, but only with less drive current (1mA). QT110H: This part is active-high, so it works in reverse to that described above.
3.2 ELECTRODE WIRING
See also Section 3.4. The wiring of the electrode and its connecting trace is important to achieving high signal levels and low noise. Certain design rules should be adhered to for best results: 1. Use a ground plane under the IC itself and Cs and Rs but NOT under Re, or under or closely around the electrode or its connecting trace. Keep ground away from these things to reduce stray loading (which will dramatically reduce sensitivity). 2. Keep Cs, Rs, and Re very close to the IC. 3. Make Re as large as possible. As a test, check to be sure that an increase of Re by 50% does not appreciably decrease sensitivity; if it does, reduce Re until the 50% test increase has a negligible effect on sensitivity. 4. Do not route the sense wire near other `live' traces containing repetitive switching signals; the sense trace will pick up noise from them.
3 - CIRCUIT GUIDELINES
3.1 SAMPLE CAPACITOR
When used for most applications, the charge sampler Cs can be virtually any plastic film or good quality ceramic capacitor. The type should be relatively stable in the anticipated temperature range. If fast temperature swings are expected, especially at higher sensitivity, a more stable capacitor might be required for example PPS film. In most moderate applications a low-cost X7R type will work fine.
3.3 POWER SUPPLY, PCB LAYOUT
See also Section 3.4. The power supply can range from 2.5 to 5.0 volts. At 2.5 volts current drain averages less than 10A with Cs = 10nF, provided a 470K Rs resistor is used (Figure 2-6). Idd curves are shown in Figure 4-4. Higher values of Cs will raise current drain. Higher Cx values can actually decrease power drain. Operation can be from batteries, but be cautious about loads causing supply droop (see Output Drive, Section 2.2.6) if the batteries are unregulated. As battery voltage sags with use or fluctuates slowly with temperature, the IC will track and compensate for these changes automatically with only minor changes in sensitivity. If the power supply is shared with another electronic system, care should be taken to assure that the supply is free of digital spikes, sags, and surges which can adversely affect the device. The IC will track slow changes in Vdd, but it can be affected by rapid voltage steps. if desired, the supply can be regulated using a conventional low current regulator, for example CMOS LDO regulators that have nanoamp quiescent currents. Care should be taken that the regulator does not have a minimum load specification, which almost certainly will be violated by the QT110's low current requirement. Furthermore, some LDO regulators are unable to provide adequate transient regulation between the
Figure 2-6 Eliminating HB Pulses
G AT E OR M IC RO INP U T
CMO S
2 Co
100p F
O UT
SN S 2
7 5 6
3 4
O PT1
GA IN
O PT2
SN S 1
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quiescent and acquire states, creating Vdd disturbances that will interfere with the acquisition process. This can usually be solved by adding a small extra load from Vdd to ground, such as 10K ohms, to provide a minimum load on the regulator. Conventional non-LDO type regulators are usually more stable than slow, low power CMOS LDO types. Consult the regulator manufacturer for recommendations. For proper operation a 100nF (0.1uF) ceramic bypass capacitor must be used between Vdd and Vss; the bypass cap should be placed very close to the device's power pins. Without this capacitor the part can break into high frequency oscillation, get physically hot, stop working, or become damaged. PCB Cleanliness: All capacitive sensors should be treated as highly sensitive circuits which can be influenced by stray conductive leakage paths. QT devices have a basic resolution in the femtofarad range; in this region, there is no such thing as `no clean flux'. Flux absorbs moisture and becomes conductive between solder joints, causing signal drift and resultant false detections or temporary loss of sensitivity. Conformal coatings will trap in existing amounts of moisture which will then become highly temperature sensitive. The designer should strongly consider ultrasonic cleaning as part of the manufacturing process, and in more extreme cases, the use of conformal coatings after cleaning and baking.
Because the charge and transfer times of the QT110 are relatively long (~2s), the circuit can tolerate a large value of Re, often more than 10k ohms in most cases.
Diodes or semiconductor transient protection devices or MOV's on the electrode trace are not advised; these devices have extremely large amounts of nonlinear parasitic capacitance which will swamp the capacitance of the electrode and cause false detections and other forms of instability. Diodes also act as RF detectors and will cause serious RF immunity problems.
3.4 EMC AND RELATED NOISE ISSUES
External AC fields (EMI) due to RF transmitters or electrical noise sources can cause false detections or unexplained shifts in sensitivity. The influence of external fields on the sensor is reduced by means of the Rseries described in Section 3.2. The Cs capacitor and Rseries (Figure 1-1) form a natural low-pass filter for incoming RF signals; the roll-off frequency of this network is defined by -
FR =
1 2R series C s
3.3.1 SUPPLY CURRENT
Measuring average power consumption is a challenging task due to the burst nature of the device's operation. Even a good quality RMS DMM will have difficulty tracking the relatively slow burst rate, and will show erratic readings. The easiest way to measure Idd is to put a very large capacitor, such as 2,700F across the power pins, and put a 220 ohm resistor from there back to the power source. Measure the voltage across the 220 resistor with a DMM and compute the current based on Ohm's law. This circuit will average out current to provide a much smoother reading. To reduce the current consumption the most, use high or low gain pin settings only, the smallest value of Cs possible that works, and a 470K resistor (Rs) across Cs (Figure 1-1). Rs acts to help discharge capacitor Cs between bursts, and its presence substantially reduces power consumption.
If for example Cs = 22nF, and Rseries = 10K ohms, the rolloff frequency to EMI is 723Hz, vastly lower than any credible external noise source (except for mains frequencies i.e. 50 / 60 Hz). However, Rseries and Cs must both be placed very close to the body of the IC so that the lead lengths between them and the IC do not form an unfiltered antenna at very high frequencies. PCB layout, grounding, and the structure of the input circuitry have a great bearing on the success of a design to withstand electromagnetic fields and be relatively noise-free. These design rules should be adhered to for best ESD and EMC results: 1. Use only SMT components. 2. Keep Cs, Rs, Re and Vdd bypass cap close to the IC. 3. Maximize Re to the limit where sensitivity is not affected. 4. Do not place the electrode or its connecting trace near other traces, or near a ground plane. 5. Do use a ground plane under and around the QT110 itself, back to the regulator and power connector (but not beyond the Cs capacitor). 6. Do not place an electrode (or its wiring) of one QT11x device near the electrode or wiring of another device, to prevent cross interference. 7. Keep the electrode (and its wiring) away from other traces carrying AC or switched signals. 8. If there are LEDs or LED wiring near the electrode or its wiring (ie for backlighting of the key), bypass the LED wiring to ground on both its ends. 9. Use a voltage regulator just for the QT110 to eliminate noise coupling from other switching sources via Vdd. Make sure the regulator's transient load stability provides for a stable voltage just before each burst commences. For further tips on construction, PCB design, and EMC issues browse the application notes and faq at www.qprox.com
3.3.2 ESD PROTECTION
In cases where the electrode is placed behind a dielectric panel, the IC will be protected from direct static discharge. However even with a panel transients can still flow into the electrode via induction, or in extreme cases via dielectric breakdown. Porous materials may allow a spark to tunnel right through the material. Testing is required to reveal any problems. The device has diode protection on its terminals which will absorb and protect the device from most ESD events; the usefulness of the internal clamping will depending on the dielectric properties, panel thickness, and rise time of the ESD transients. The best method available to suppress ESD and RFI is to insert a series resistor Re in series with the electrode as shown in Figure 1-1. The value should be the largest that does not affect sensing performance. If Re is too high, the gain of the sensor will decrease.
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QT110/110H R1.03/0604
4.1 ABSOLUTE MAXIMUM SPECIFICATIONS
Operating temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix Storage temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -55OC to +125OC VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6.5V Max continuous pin current, any control or drive pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20mA Short circuit duration to ground, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite Short circuit duration to VDD, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite Voltage forced onto any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.6V to (Vdd + 0.6) Volts
4.2 RECOMMENDED OPERATING CONDITIONS
VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.5 to 5.5V Supply ripple+noise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10mV p-p max Max Cx load capacitance. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20pF Cs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10nF ~ 22nF X7R ceramic
Rs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 470K 5%
4.3 AC SPECIFICATIONS
Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Ta = 20OC, unless otherwise noted. Parameter
TRC TQ TBS TBL TR FP TP TPO THB FQ
Description
Recalibration time Charge, transfer duration Burst spacing interval Burst length Response time Piezo drive frequency Piezo drive duration Pulse output width on Out Heartbeat pulse width Burst frequency
Min
Typ
550 2 75 95
Max
Units
ms s ms ms @ 5.0V Vdd @ 3.3V Vdd
Notes
0.5 129 3.6 4 75 75 300 165
7 4.4
ms ms kHz ms ms s kHz
4.4 SIGNAL PROCESSING
Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Ta = 20OC, unless otherwise noted. Description
Threshold differential, high gain Threshold differential, medium gain Threshold differential, low gain Hysteresis Consensus filter length Positive drift compensation rate Negative drift compensation rate Post-detection recalibration timer duration 10
Min
Typ
3.1 4.7 6.25 50 4 750 75
Max
Units
% % % % samples ms/level ms/level Note 1 Note 1 Note 1 Note 2
Notes
60
secs
Note 3
Note 1: Of absolute full scale signal Note 2: Of signal threshold Note 3: Strap option.
LQ
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QT110/110H R1.03/0604
4.5 DC SPECIFICATIONS
Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20OC, unless otherwise noted. Parameter Description Min Typ Max Units
VDDL IDD Guaranteed min Vdd Supply current 26 12 9.5 100 0.8 2.2 0.6 Vdd-0.7 1 8 1.2 1.8 3.8 2.45 V A A A V/s V V V V A bits pF pF pF Cx = 10pF, Cs = 15nF; Figure 4-1 Cx = 10pF, Cs = 15nF; Figure 4-2 Cx = 10pF, Cs = 15nF; Figure 4-3 @5.0V @3.3V @2.5V Required for proper startup OPT1, OPT2 OPT1, OPT2 OUT, 4mA sink OUT, 1mA source OPT1, OPT2
Notes
VDDS VIL VHL VOL VOH IIL AR S[1] S[2] S[3]
Supply turn-on slope Low input logic level High input logic level Low output voltage High output voltage Input leakage current Acquisition resolution Sensitivity - high gain Sensitivity - medium gain Sensitivity - low gain
Figure 4-1 High Gain Sensitivity and Range @ Vdd = 3V
3.0 4.0
Figure 4-2 Medium Gain Sensitivity and Range @ Vdd = 3V
Sensitivity, p F
2.0 1.5 1.0 0.5
25pF 20pF 10pF 5pF 0pF Valid operating range
10 20 30
Sensitivity, p F
2.5
Cx=30pF
3.0
Cx=30pF 25pF 20pF
2.0
10pF 5pF 0pF Valid operating range
10 20 30
1.0
Cs, nF
Figure 4-3 Low Gain Sensitivity and Range @ Vdd = 3V
40
Cs, nF
Figure 4-4 Typical Supply Current Vs Vdd Rs = 470K, Cx = 10pF, Gain = High
8.0
Idd, Microamperes
Cx=30pF 25pF 20pF
35 30 25 20 15 10 5 Cs = 10nF Cs = 20nF
Sensitivity, pF
6.0
4.0
10pF 5pF 0pF Valid operating range
10 20 30
. ..
2.0
2.5
3
3.5
4
4.5
5
Cs, nF
Vdd
LQ
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QT110/110H R1.03/0604
4.6 MECHANICAL
Package type: 8pin Dual-In-Line
SYMBOL
a A M m Q P L L1 F R r S S1 Aa x Y
Min
6.096 7.62 9.017 7.62 0.889 0.254 0.355 1.397 2.489 3.048 0.381 3.048 7.62 8.128 0.203
Millimeters Max
7.112 8.255 10.922 7.62 0.559 1.651 2.591 3.81 3.556 4.064 7.062 9.906 0.381
Notes
Min
0.24 0.3 0.355 0.3 0.035 0.01 0.014 0.055 0.098 0.12 0.015 0.12 0.3 0.32 0.008
Inches Max
0.28 0.325 0.43 0.3 0.022 0.065 0.102 0.15 0.14 0.16 0.3 0.39 0.015
Notes
Typical BSC
Typical BSC
Typical
Typical
BSC
BSC
Package type: 8pin SOIC
SYMBOL
M W Aa H h D L E e O
Min
4.800 5.816 3.81 1.371 0.101 1.27 0.355 0.508 0.19 0.381 0
Millimeters Max
4.979 6.198 3.988 1.728 0.762 1.27 0.483 1.016 0.249 0.762 8
Notes
Min
0.189 0.229 0.15 0.054 0.004 0.050 0.014 0.02 0.007 0.229 0
Inches Max
0.196 0.244 0.157 0.068 0.01 0.05 0.019 0.04 0.01 0.03 8
Notes
BSC
BSC
LQ
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QT110/110H R1.03/0604
5 - ORDERING INFORMATION
PART
QT110-D QT110-IS QT110H-D QT110H-IS
TEMP RANGE
0 - 70C -40 - 85C 0 - 70C -40 - 85C
PACKAGE
PDIP SOIC-8 PDIP SOIC-8
MARKING
QT1 + 10 or QT110 QT1 + I or QT110-I QT1 +10H or QT110H QT1 + AI or QT110H-I
LQ
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QT110/110H R1.03/0604
lQ
Copyright (c) 1999 - 2004 QRG Ltd. All rights reserved. Patented and patents pending
Corporate Headquarters
1 Mitchell Point Ensign Way, Hamble SO31 4RF Great Britain Tel: +44 (0)23 8056 5600 Fax: +44 (0)23 80565600 admin@qprox.com
www.qprox.com North America
651 Holiday Drive Bldg. 5 / 300 Pittsburgh, PA 15220 USA Tel: 412-391-7367 Fax: 412-291-1015
The specifications set out in this document are subject to change without notice. All products sold and services supplied by QRG are subject to our Terms and Conditions of sale and supply of services which are available online at www.qprox.com and are supplied with every order acknowledgment. QProx, QTouch, QMatrix, QWheel, QLevel, and QSlide are trademarks of QRG. QRG products are not suitable for medical (including lifesaving equipment), safety or mission critical applications or other similar purposes. Except as expressly set out in QRG's Terms and Conditions, no licenses to patents or other intellectual property of QRG (express or implied) are granted by QRG in connection with the sale of QRG products or provision of QRG services. QRG will not be liable for customer product design and customers are entirely responsible for their products and applications which incorporate QRG's products.

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